Transmitter with internal compensation for variance in differential data line impedance

ABSTRACT

In at least some embodiments, an electronic device includes a first data endpoint and differential data transceiver coupled to the first data endpoint. The differential transceiver provides a communication interface between the first data endpoint and a second data endpoint. The differential transceiver compensates for variations in a series impedance and/or a parallel impedance for a differential data line between the differential transceiver and the second data endpoint.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a non-provisional application claiming priority toEuropean Patent Application Serial No. EP 09290881.3, filed on Nov. 25,2009, and entitled “Transmitter With Internal Compensation For VarianceIn Differential Data Line Series Impedance”, the teachings of which areincorporated by reference herein.

BACKGROUND

In order for high-speed differential data communications to besuccessful, predetermined communication parameters need to be met. Forexample, the transmitter output impedance, the data line characteristicimpedance, and the receiver input impedance should be matched to limitdata reflections.

SUMMARY

In at least some embodiments, an electronic device comprises a firstdata endpoint and differential data transceiver coupled to the firstdata endpoint. The differential data transceiver providing acommunication interface between the first data endpoint and a seconddata endpoint. The differential transceiver internally compensates forvariations in a series impedance for a differential data line betweenthe differential transceiver and the second data endpoint.

In at least some embodiments, a differential data transmitter comprisesa differential data line port and programmable circuitry coupled to thedifferential data line port. The programmable circuitry internallycompensates for external variances in a differential data line seriesimpedance.

In at least some embodiments, a method is performed for a differentialdata transmitter. The method comprises detecting a variance in adifferential data line series impedance and programming the differentialdata transmitter to internally compensate for the variance.

BRIEF DESCRIPTION OF THE DRAWINGS

For a detailed description of various embodiments of the invention,reference will now be made to the accompanying drawings in which:

FIG. 1 shows a system in accordance with an embodiment of thedisclosure;

FIG. 2 shows an embodiment of the programmable circuitry described forFIG. 1;

FIG. 3 shows a printed circuit board of an electronic device inaccordance with an embodiment of the disclosure;

FIG. 4 illustrates waveform variations output by a differential datatransmitter in accordance with an embodiment of the disclosure; and

FIG. 5 shows a method in accordance with an embodiment of thedisclosure.

NOTATION AND NOMENCLATURE

Certain terms are used throughout the following description and claimsto refer to particular system components. As one skilled in the art willappreciate, computer companies may refer to a component by differentnames. This document doe not intend to distinguish between componentsthat differ in name but not function. In the following discussion and inthe claims, the terms “including” and “comprising” are used in anopen-ended fashion, and thus should be interpreted to mean “including,but not limited to . . . .” Also, the term “couple” or “couples” isintended to mean either an indirect or direct electrical connection.Thus, if a first device couples to a second device, that connection maybe through a direct electrical connection, or through an indirectelectrical connection via other devices and connections. The term“system” refers to a collection of two or more hardware and/or softwarecomponents, and may be used to refer to an electronic device or devicesor a sub-system thereof.

DETAILED DESCRIPTION

While the invention is susceptible to various modifications andalternative forms, specific embodiments thereof are shown by way ofexample in the drawings and will herein be described in detail. Itshould be understood, however, that the drawings and detaileddescription thereto are not intended to limit the invention to theparticular form disclosed, but on the contrary, the intention is tocover all modifications, equivalents and alternatives falling within thespirit and scope of the present invention as defined by the appendedclaims.

Embodiments of the invention are directed to differential datatransmitters and related systems and methods. In at least someembodiments, a differential data transmitter internally compensates forexternal variations in the series impedance of a differential data linecoupled thereto. In this manner, predetermined requirements fordifferential data communications can be met while maintainingflexibility with external device components or circuitry that aredesired or inherent for different product designs, but that affect theseries impedance of the differential data line.

FIG. 1 shows a system 100 in accordance with an embodiment of thedisclosure. As shown, the system 100 comprises a first electronic device102 and a second electronic device 160 configured to communicate via adifferential data line 130. More specifically, the first electronicdevice 102 comprises a differential data transceiver 104 coupled to adata endpoint 106. Similarly, the second electronic device 160 comprisesa differential data transceiver 140 coupled to a data endpoint 150. Eachof the data endpoints 106, 150 corresponds to an addressable componentthat is the source or sink of information transmitted between the firstand second electronic devices 102, 160. As an example, each of theelectronic devices 102, 160 may be a host device or a peripheral devicecompatible with the Universal Serial Bus (USB) 2.0 protocol or anotherdifferential data communication protocol.

As shown, the differential data transceiver 104 comprises a high-speedreceiver 110 and a high-speed transmitter 108 coupled between the dataendpoint 106 and the differential data line 130. In operation, thehigh-speed transmitter 108 receives data from the data endpoint 106 andcontrols the voltage levels on the differential data line 130 totransmit a representation of this data over the different data line 130.Meanwhile, the high-speed receiver 110 is configured to examine thevoltage levels on the different data line 130 and to decode dataaccordingly. The decoded data is then forwarded to the data endpoint106. If the first electronic device 102 has multiple data endpoints, asmay be the case in some embodiments, the high-speed receiver 110 is beto forward the data to the correct data endpoint based on an addressingscheme. The operation of the differential data transceiver 140 and itscomponents (e.g., the high-speed transmitter 144 and the high-speedreceiver 142) can be understood to be similar to the operation of thedifferential data transceiver 104 and its components. However, at leastone of the differential data transceivers 104, 160 comprisesprogrammable circuitry that enables compensation for external variationsin the series impedance of the differential data line 130. In FIG. 1,the differential data transceiver 104 is shown to have this programmablecircuitry 120. However, such programmable circuit could additionally oralternatively be implemented with the differential data transceiver 140.

In accordance with at least some embodiments, the programmable circuitry120 compensates for variations in the series impedance of thedifferential data line 130 by programming the resistance of single-endedtermination resistors (Z_(TERM)) 122 for the differential data line 130.Additionally or alternatively, the programmable circuitry 120compensates for variations in the series impedance by varying aprogrammable transmitter output current drive. Additionally oralternatively, the programmable circuitry 120 compensates for variationsin the series impedance by varying a programmable transient boostcurrent. Additionally or alternatively, the programmable circuitry 120compensates for variations in the parallel impedance by varying aprogrammable transient boost current. In accordance with at least someembodiments, the transient boost current opens an eye diagram duringdata transitions without violating communication protocol requirementsfor direct current (DC) values. Although FIG. 1 shows the programmablecircuitry 120 and termination resistors 122 as being separate from thehigh-speed transmitter 108 (between the transmitter 108 and thedifferential data line 130), alternative embodiments may combine thesecomponents together. Additional details for the programmable circuitry120 are provided in FIG. 2.

FIG. 2 shows an embodiment of the programmable circuitry or impedancecompensation circuitry 120 described for FIG. 1. In the embodiment ofFIG. 2, the programmable circuitry 120 generally comprises a firstcircuitry portion or reference current source 250, a second circuitryportion or programmable current source 202, a third circuitry portion ora control circuit 207, a fourth circuitry portion or a current mirror220, and a fifth circuitry portion or boost current circuit 230. As willbe described in greater detail, the first circuitry portion 250, thesecond circuitry portion 202, and the third circuitry portion 207 enabletermination (ZHS) impedances 214A and 214B to be controlled. Firstly,the first, second and third circuitry portions allow ZHS impedances tobe compensated for temperature drift and therefore maintain impedancematching with the external termination impedances ZTERM 280. Secondly,the first, second and third circuitry portions allow ZHS impedances tobe programmed to compensate for external series impedances (ZSER) 260added to the differential data line 130 between the circuitry 120 andthe connector 270.

As a first example, the programmable circuitry 120 may be used tocalibrate termination (ZHS) impedances 214A and 214B to match with theexternal termination impedances ZTERM 280 such that ZHS=ZTERM. As asecond example, the programmable circuitry 120 may be used to programtermination (ZHS) impedances 214A and 214B to compensate for an externalseries impedances (ZSER) 260 while maintaining matching with theexternal termination impedances ZTERM 280 at the connector 270 such thatZHS=ZTERM−ZSER.

As shown, the first circuitry portion 250 provides a current sourceIHSREF which should be first-order temperature independent. Oneimplementation of such a current source is shown in FIG. 2. In thisexample, a bandgap reference voltage is applied to a resistive element258 via a feedback loop comprising an amplifier 252 and a transistor256. The resistive element 258 may take the form of, but is not limitedto, an internal resistor, an external resistor, or a switched capacitorresistive element. The second circuitry portion 202 may comprisetransistors 204 and 206. The transistor 204 controls a programmablecurrent I_(REP) based on IHSREF and a control signal (ZREP_PROG). Thetransistor 206 controls a fixed current I_(C1) based on IHSREF. Inaccordance with at least some embodiments, the transistors 204 representa current digital-to-analog converter (DAC) (e.g., a 3-bit current DAC)with ZREP_PROG as the control signal. ZREP_PROG control signal isprovided externally.

As shown, I_(REP) is provided from the second circuitry portion 202 tothe third circuitry portion 207. In accordance with at least someembodiments, the second circuitry portion comprises a finite statemachine (FSM) 208 that provides a first control signal (R_(TERM) _(—)_(CAL)) for a variable resistor (Z_(REP)) and a second control signal(R_(TERM) _(—) _(RMX)) for the variable ZHS impedances 214A and 214B.The operation of the FSM 208 is controlled in part by the output of acomparator 210 that compares an input signal (V_(REP)) with a voltagereference (VHSREF).

In accordance with at least some embodiments, the ZHS impedances 214A,214B represent a resistive digital-to-analog converter (DAC) or otherprogrammable component with the input control code being provided by theFSM 208. To calibrate the ZHS impedances 214A and 214B, the thirdcircuitry portion 207 is used. A replica (I_(REP)) of the temperatureindependent output drive current (IHS) is applied through a replica(Z_(REP)) of the temperature dependent ZHS impedances 214A, 214B. Thecalibration of Z_(REP) generates a temperature dependent voltage(V_(REP)) which is a proportional replica of the high level transmitteroutput voltage (VHSOH_conn).

VHSOH_ic is the transmitter output voltage of the transceiver 104 at theintegrated circuit (IC) pins (pins DP and DM in FIG. 2).VHSOH_ic=IHS*(ZHS//((ZSER+ZTERM))=IHS*(ZHS*(ZSER+ZTERM)/(ZSER+ZHS+ZTERM)),where “//” denotes “in parallel with”. VHSOH_conn is the transmitteroutput voltage of the transceiver 104 measured at the connector 270.Hence VSHOH_conn=VHSOH_ic*(ZTERM/(ZSER+ZTERM))=IHS*ZHS*ZTERM/(ZSER+ZHS+ZTERM).

As shown ZHS=ZTERM−ZSER is required in order to compensate for anexternal series impedances (ZSER) 260 while maintaining matching withthe external termination impedances ZTERM 280 at the connector 270.Substituting for this ZHS target in the VHSOH_conn equation results inVHSOH_conn=IHS*(ZTERM−ZSER)/2=IHS*ZHS/2.

V_(REP) is compared to a temperature independent voltage reference(VHSREF) by the comparator 210 and the comparator's output is providedas a control signal to the FSM 208. The FSM 208 then outputs a controlsignal (e.g., a control code) to drive the termination ZHS impedances214A and 214B up or down.

In FIG. 2, the second circuitry portion 202, the fourth circuitryportion 220, and the fifth circuitry portion 230 combined show that fora given IHS_PROG multiplication factor M (described later), IHS isrelated to I_(REP) by a fixed current multiplication ratio Ki, where Kiis defined by the equation I_(REP)=Ki*IHS. A fixed ratio between Z_(REP)and ZHS is maintained irrespective of the calibrated value of Z_(REP),such that we can define Z_(REP)=Kr*ZHS. If a resistive DAC is used forthe ZHS impedances 214A and 214B, and for the replica (Z_(REP))impedance 212, the FSM 208 may sweep down a sequence of input codes forthe replica resistive DAC Z_(REP) 212 until the comparator 210 togglesfrom high to low. This sets the value of Z_(REP) 212 and hence ZHS. Inaccordance with embodiments, the toggling occurs approximately whenVHSREF=V_(REP). Since V_(REP)=I_(REP)*Z_(REP); and I_(REP)=Ki*IHS; andZ_(REP)=Kr*ZHS; and VHSOH_conn=IHS*(ZHS/2)); hence,V_(REP)=2*Ki*Kr*VHSOH_conn. Since Ki and Kr are temperature independentfactors and VHSREF is a bandgap voltage and is (to first-orderapproximation) temperature independent, VHSOH_conn can be madetemperature independent. Thus, after calibration we haveV_(REP)=VHSREF=I_(REP)*Z_(REP). Substituting for Z_(REP)=Kr*ZHS andsolving for ZHS, we have ZHS=(VHSREF/I_(REP))*(1/Kr). Therefore 3independent variables are available with which to calibrate and programZHS. By programming I_(REP), the ZHS impedances 214A and 214B can bemade programmable.

In accordance with at least some embodiments, I_(REP) is programmedusing an offset current DAC (implemented by the second circuitry portion202). For example, a DAC of 3-bits may be controlled by a digital signalZREP_PROG allowing IREP to be independently adjusted by a multiplicationfactor P. In this case, the equation for ZHS becomesZHS=(VHSREF/(I_(REP)*P)*(1/Kr). Using this multiplication factor P, theZHS impedances 214A, 214B can be reduced to compensate for seriesimpedance on the differential data line 130. If VHSREF, Kr and thedefault value of I_(REP) are set in order to calibrate for ZHS=ZTERMwith multiplication factor P=1 (where P corresponds to a particularvalue of ZREP_PROG); then P can be programmed to compensate for externalseries impedances (ZSER) 260 by adjusting the calibration target toZTERM−ZSER by setting P=ZTERM/(ZTERM−ZSER).

As an example, a system with termination impedances ZTERM 280 may havevarious external components (e.g., common mode filters), which may becoupled to the differential data line 130 and which present a combinedseries impedance ZSER 260. ZSER may be compensated by programmingI_(REP) in order to have ZHS=ZTERM−ZSER. A ZHS impedance reductionbetween 0 to N ohms (let N=3 in this example) may be sufficient tocompensate for various external components (e.g., common mode filters).More specifically, if an external component adds approximately 0 ohms tothe series impedance of the differential data line 130 which isterminated with termination impedances ZTERM 280 of 45 ohms, I_(REP) isprogrammed so that each ZHS impedance 214A, 214B has a value ofapproximately 45 ohms. If an external component (or components) addsapproximately 1 ohm to the series impedance of the differential dataline 130, I_(REP) is programmed so that each ZHS impedance 214A, 214Bhas a value of approximately 44 ohms. If an external component (orcomponents) adds approximately 2 ohms to the series impedance of thedifferential data line 130, I_(REP) is programmed so that each ZHSimpedance 214A, 214B has a value of approximately 43 ohms. If anexternal component (or components) adds approximately 3 ohms to theseries impedance of the differential data line 130, I_(REP) isprogrammed so that each ZHS impedance 214A, 214B has a value ofapproximately 42 ohms.

When an external component (or components) is added to the differentialdata line 130, the high-level voltage at the transceiver 104 (VHSOH_ic)is not the same as the high-level voltage at connector 270 (VHSOH_conn)because VHSOH_conn is degraded by the serial impedance ZSER 260 on thedifferential data line 130. This may immediately be seen from theequation for VSHOH_conn described previously by setting ZHS=ZTERM−ZSER.The result is VHSOH_conn=IHS*(ZTERM−ZSER)/2, which is less than thetarget value of VHSOH=VHSOH_ic=VHSOH_conn=IHS*ZTERM/2 for matched linewith ZSER=0 and ZHS=ZTERM. This degradation in VHSOH_conn may becompensated for by multiplying by an independent variable M such thatVHSOH_conn=M*IHS*(ZTERM−ZSER)/2. For the case of no external seriesimpedance present (i.e., ZSER=0), setting M=1 will makeVHSOH_conn=IHS*ZTERM/2. For the case of an external series impedanceZSER present, setting M=(ZTERM/(ZTERM−ZSER)) will makeVHSOH_conn=IHS*ZTERM/2.

In accordance with at least some embodiments, the fourth circuitryportion 220 is configured to adjust IHS by a multiplication factor M. Asshown, the fourth circuitry portion 220 comprises transistors 222 and224, which may represent an offset current DAC (e.g., a 4-bit DAC) oranother programmable component controlled by a digital control signalIHS_PROG in order to set the IHS multiplication factor M. The IHS_PROGcontrol signal may be provided externally. Increasing and decreasing IHShas a direct effect on the VHSOH_conn values. In other words, increasingIHS causes VHSOH_conn to increase. Alternatively, decreasing IHS causesVHSOH_conn to decrease. In accordance with at least some embodiments,the IHS values and the ZHS impedance values are controlled together. Aspreviously discussed, the ZHS impedances 214A, 214B are programmable bycontrolling I_(REP) (e.g., using the first circuitry portion 250, thesecond circuitry portion 202, and the third circuitry portion 207) andVHSOH is programmable by controlling IHS (e.g., using the fourthcircuitry portion 220). As previously discussed, in some embodiments,I_(REP) and IHS may be controlled using current DACs.

As an example Table 1A shows various parameter values and illustrateshow VHSOH_conn values are affected by ZSER 260; Table 1B shows variousparameter values and illustrates how VHSOH_conn values are affected byadjusting the ZHS impedances 214A, 214B to compensate for ZSER 260; andTable 1C shows various parameter values and illustrates how VHSOH_connvalues are affected by adjusting the ZHS impedances 214A, 214B and atransmitter output current drive (IHS) multiplication factor M tocompensate for ZSER 260. In Tables 1A-1C, it is assumed that the desiredvoltage level for VHSOH_conn is 400.0 mV with termination impedancesZTERM 280 of 45 ohms. For example, a particular communication protocol(e.g., USB 2.0) may specify that voltage levels for differential datacommunications be at 400 mV+/−10% (360 mV to 440 mV).

TABLE 1A ZSER (ohm) 0 1 2 3 IHS (mA) 17.78 17.78 17.78 17.78 M 1.0001.000 1.000 1.000 ZHS (ohm) 45 45 45 45 ZTERM (ohm) 45 45 45 45 VHSOH_ic(mV) 400.0 404.4 408.7 412.9 VHSOH_conn (mV) 400.0 395.6 391.3 387.1

TABLE 1B ZSER (ohm) 0 1 2 3 IHS (mA) 17.78 17.78 17.78 17.78 M 1.0001.000 1.000 1.000 ZHS (ohm) 45 44 43 42 ZTERM (ohm) 45 45 45 45 VHSOH_ic(mV) 400.0 400.0 400.0 400.0 VHSOH_conn (mV) 400.0 391.3 383.0 375.0

TABLE 1C ZSER (ohm) 0 1 2 3 IHS (mA) 17.78 18.17 18.57 18.96 M 1.0001.022 1.044 1.066 ZHS (ohm) 45 44 43 42 ZTERM (ohm) 45 45 45 45 VHSOH_ic(mV) 400.0 408.9 417.8 426.7 VHSOH_conn (mV) 400.0 400.0 400.0 400.0

As shown in the example of Table 1A, VHSOH_conn decreases as ZSER 260increases. More specifically, if ZSER is approximately 0 ohms,VHSOH_conn is approximately 400.0 mV. If ZSER is approximately 1 ohm,VHSOH_conn is approximately 395.6 mV. If ZSER is approximately 2 ohms,VHSOH_conn is approximately 391.3 mV. If ZSER is approximately 3 ohms,VHSOH_conn is approximately 387.1 mV. In comparison, VHSOH_ic increasesfrom 400.0 mV to 412.9 mV as ZSER increases from 0 ohms to 3 ohms.Various other parameters shown for FIG. 2 maintain their respectivevalues as ZSER changes, with IHS at 17.78 mA, M at 1, ZHS at 45 ohms,and ZTERM at 45 ohms.

As shown in the example of Table 1B, adjusting the ZHS impedances 214A,214B to compensate for ZSER 260 sets VHSOH_ic at a desired value (e.g.,400.0 mV), but causes an additional decrease in VHSOH_conn values. Morespecifically, if ZSER is approximately 0 ohms, ZHS is approximately 45ohms and VHSOH_conn is approximately 400.0 mV. If ZSER is approximately1 ohm, ZHS is reduced to approximately 44 ohms and VHSOH_conn isapproximately 391.3 mV. If ZSER is approximately 2 ohm, ZHS is reducedto approximately 43 ohms and VHSOH_conn is approximately 383.0 mV. IfZSER is approximately 3 ohm, ZHS is reduced to approximately 42 ohms andVHSOH_conn is approximately 375.0 mV. Various other parameters shown forFIG. 2 maintain their respective values as ZSER changes, with IHS at17.78 mA, M at 1, and ZTERM at 45 ohms.

As shown in the example of Table 1C, adjusting the ZHS impedances 214A,214B and IHS multiplication factor M together to compensate for ZSER 260increases VHSOH_conn values (to be nearer a desired value such as 400.0mV) compared to adjusting only ZHS impedances 214A, 214B as in Table 1B.More specifically, if ZSER is approximately 0 ohms, ZHS is approximately45 ohms, IHS is approximately 17.78 mA, M is approximately 1, VHSOH_icis approximately 400.0 mV, and VHSOH_conn is approximately 400.0 mV. IfZSER is approximately 1 ohm, ZHS is adjusted to approximately 44 ohms,IHS is adjusted to approximately 18.17 mA with M approximately 1.022,VHSOH_ic is adjusted to approximately 408.9 mV, and VHSOH_conn isapproximately 400.0 mV. If ZSER is approximately 2 ohms, ZHS is adjustedto approximately 43 ohms, IHS is adjusted to approximately 18.57 mA withM approximately 1.044, VHSOH_ic is adjusted to approximately 417.8 mV,and VHSOH_conn is approximately 400.0 mV. If ZSER is approximately 3ohms, ZHS is adjusted to approximately 42 ohms, IHS is adjusted toapproximately 18.96 mA with M approximately 1.066, VHSOH_ic is adjustedto approximately 426.7 mV, and VHSOH_conn is approximately 400.0 mV.Other parameters shown for FIG. 2 maintain their respective values asZSER, IHS, and M change, with ZTERM at 45 ohms.

In accordance with at least some embodiments, a transient boost currentmay also be added to IHS. In FIG. 2, the transient boost current iscontrolled by the fifth circuitry portion 230, which selectivelyprovides the boost current during voltage transitions (from low-to-highor from high-to-low). As shown, the fifth circuitry portion 230comprises a current minor (which generally comprises transistors 236 and238), a boost transistor 234, a transition detector 232, and a switchnetwork 240. The current minor (which generally comprises transistors236 and 238) mirrors a current Ic2 controlled by the fourth circuitryportion 220 to provide the transmitter output drive current IHS and toselectively boost IHS (e.g., during transitions), and the transistor 234acts as an auxiliary transmitter to selectively boost IHS based on acontrol signal from transition detector 232. The current mirror (i.e.,transistors 236 and 238) and boost transistor 234 can be coupled to thedifferential terminals DP and DM through the switch network 240 when avoltage transition is detected by the transition detector 232. As anexample, the boost may add approximately 10% of extra current to IHS. Inaccordance with at least some embodiments, the transition detector 232is reset after the output data is stable for a predetermined number ofconsecutive periods (e.g., 2.5 periods) and the current boost stops atleast until a subsequent transition is detected.

FIG. 3 shows a printed circuit board (PCB) 300 of an electronic device(e.g., device 102) in accordance with an embodiment of the disclosure.In FIG. 3, the PCB 300 comprises a differential data transceiver 302mounted on the PCB 300, the differential data transceiver 302 representsthe transceiver 104 of FIG. 1 or another transceiver that implements theprogrammable circuitry 120 previously discussed. As shown, the PCB 300comprises various components external to the transceiver 302, includinga PCB trace 304, a PCB series element 306 (i.e., the PCB series element306 is in series with the PCB trace 304 which forms part of adifferential data line), a PCB parallel element 308 (i.e., the PCBparallel element 308 is in parallel with the PCB trace 304 which formspart of a differential data line), and a PCB connector 310. The PCBseries element 306 may represent, for example, a common mode filter. ThePCB parallel element 308 may represent, for example, an electrostaticdischarge (ESD) protection element. The PCB connector 310 is compatible,for example, with USB 2.0 protocol or another communication protocol.The PCB trace 304, the PCB series element 306, and the PCB connector 310may add series impedance to the differential data line associated withthe transceiver 302. To compensate for such series impedance on thedifferential data line, the programmable circuitry 120 of transceiver302 is configured to adjust ZHS, the transmitter output drive currentIHS and/or the transient current boost as previously discussed. Tocompensate for parallel impedance on the differential line, which maycause transient reflections on the line due to discontinuities in lineimpedance matching, or other transient reflections, the transmitteroutput drive current IHS and/or the transient current boost as may beused.

FIG. 4 illustrates waveform variations 404A, 404B, 404C output by adifferential data transmitter (e.g., transmitter 104) in accordance withan embodiment of the disclosure. In FIG. 4, it is assumed that the ZHSimpedances 214A, 214B have been reduced, if necessary, to compensate forserial impedance on the differential data line. Further, it is assumedthat to comply with a communication protocol, the transmitter shouldoutput differential data DC voltage levels of 400 mV+/−10% (360 mV to440 mV).

In FIG. 4, the waveform 404A represents the output of a differentialdata transmitter without current drive compensation (IHS adjustment).During transitions (from low-to-high or from high-to-low), the waveform404A does not adequately reach +/−360 mV until some time into a firstpost-transition period 402A. The waveform 404B represents the output ofa differential data transmitter with current drive compensation (IHSadjustment). During transitions, the waveform 404B adequately reaches+/−360 mV before the first post-transition period 402A but then fallsbelow 360 mV during the first post-transition period 402A due totransient reflections on the differential data line. The waveform 404Crepresents the output of a differential data transmitter with currentdrive compensation (IHS adjustment) and current boost. Duringtransitions, the waveform 404D adequately reaches +/−360 mV before thefirst post-transition period 402A and maintains a value between 360 mVand 440 mV as is desired for the embodiment shown. After the firstpost-transition period 402A and a second post-transition period 402B,the current boost is turned off with the waveform maintaining a stablevalue at around +/−400 mV. Although FIG. 4 illustrates that currentdrive compensation and current boost compensation may be beneficial tomeet a given communication protocol requirement, different embodimentsmay differ with regard to the amount or use of ZHS impedance adjustment,current drive adjustment, and current boost adjustment.

FIG. 5 shows a method 500 in accordance with an embodiment of thedisclosure. In accordance with embodiments, the method 500 is performedfor a differential data transmitter. In FIG. 5, the method 500 comprisesdetecting a variance (from a desired value) in a differential data lineseries impedance (block 502). The detection process may be based onelectrical tests and/or physical inspection of PCB components orcomponent specifications. At block 504, the method 500 comprisesprogramming the differential data transmitter to internally compensatefor the detected variance.

In at least some embodiments, internally compensating for the variance(as in block 504) comprises adjusting a single-ended terminationresistance (ZHS). For example, in some embodiments, the single-endedtermination resistance is adjusting by providing a multi-bit controlsignal to a resistive digital-to-analog converter (DAC). Additionally,internally compensating for the variance (as in block 504) may compriseadjusting a transmitter output current drive. For example, in someembodiments, changing the transmitter output current drive comprisesproviding a multi-bit control signal to a current digital-to-analogconverter (DAC). Additionally, internally compensating for the variance(as in block 504) may comprise detecting a data transition andactivating a transient boost current.

In accordance with embodiments, the method 500 is performed when a chiphaving the differential data transmitter is incorporated into a productdesign. Thus, a single differential data transmitter design (embodied inthe chip) is compatible with many product designs, even if the productdesign couples different components to the differential data line orotherwise causes variance to the differential data line seriesimpedance.

The above discussion is meant to be illustrative of the principles andvarious embodiments of the present invention. Numerous other variationsand modifications will become apparent to those skilled in the art oncethe above disclosure is fully appreciated. It is intended that thefollowing claims be interpreted to embrace all such variations andmodifications.

1. An apparatus comprising: a first data endpoint; and differential datatransceiver coupled to the first data endpoint, the differentialtransceiver providing a communication interface between the first dataendpoint and a second data endpoint, wherein the differential datatransceiver internally compensates for variations in a series impedancefor a differential data line between the differential transceiver andthe second data endpoint based at least on part on a programmabletransient boost current, wherein the programmable transient boostcurrent opens an eye diagram during data transitions without affecting adirect current (DC) value for the data transitions.
 2. The apparatus ofclaim 1, wherein the differential data transceiver internallycompensates for variations in the series impedance based on aprogrammable single-ended termination resistance.
 3. The apparatus ofclaim 1, wherein the differential data transceiver internallycompensates for variations in the series impedance based on aprogrammable transmitter output current drive.
 4. The apparatus of claim1, wherein the apparatus further comprises a printed circuit board (PCB)upon which the differential data transceiver is mounted, whereinvariations in the series impedance for the differential data line aredue to at least one component associated with the PCB.
 5. The apparatusof claim 4, wherein the at least one component further comprises acomponent selected from the list consisting of a PCB trace, a PCBcomponent mounted in series with the differential data line, a PCBcomponent mounted in parallel with the differential data line, PCBconnector for the second data endnode.
 6. An apparatus comprising: adifferential data line port; programmable circuitry coupled to thedifferential data line port, wherein the programmable circuitryinternally compensates for external variances in a differential dataline series impedance; and a transition detector coupled to a transientcurrent booster that selectively increases an output current for thedifferent data line.
 7. The apparatus of claim 6, wherein theprogrammable circuitry comprises a resistive digital-to-analog converter(DAC) that selectively offsets a termination impedance of thedifferential data line.
 8. The apparatus of claim 6, wherein theprogrammable circuitry compensates for external variances in adifferential data line parallel impedance using at least one of atransient current boost and a programmable transmitter output currentdrive.
 9. The apparatus of claim 6, wherein the programmable circuitrycomprises a current DAC that selectively offsets a transmitter outputcurrent drive.
 10. The apparatus of claim 6, wherein the transientcurrent booster is limited to avoid surpassing a direct current (DC)value for the data transitions and wherein the transition detector isreset when output data is stable for a predetermined number of clockperiods.
 11. A method comprising: detecting a variance in a differentialdata line series impedance or a differential data line parallelimpedance; and programming the differential data transmitter tointernally compensate for the variance by configuring data transitioncircuitry to activate and de-activate a transient boost current.
 12. Themethod of claim 11, wherein the step of programming the differentialdata transmitter to internally compensate variance in the differentialdata line series impedance further comprises adjusting a single-endedtermination resistance.
 13. The method of claim 12, wherein the step ofadjusting the single-ended termination resistance further comprisesproviding a multi-bit control signal to a resistive DAC.
 14. The methodof claim 11, wherein the step of programming the differential datatransmitter to internally compensate for the variance further comprisesadjusting a transmitter output current drive.
 15. The method of claim 14wherein adjusting the transmitter output current drive comprisesproviding a multi-bit control signal to a current digital-to-analogconverter (DAC).
 16. An apparatus comprising: a pair of differentialterminals; a receiver that is coupled to the pair of differentialterminals; a transmitter that is coupled to the pair of differentialterminals; impedance compensation circuitry including: a control circuitthat is coupled to the pair of differential terminals so as to provide atemperature dependant termination impedance; and a boost current circuitthat is coupled to the pair of differential terminals so as to provide aboost current during voltage transitions.
 17. The apparatus of claim 16,wherein the impedance compensation circuitry further comprises: areference current source; a programmable current source that is coupledto the reference current source and the control circuit, wherein theprogrammable current source provides a first current to the controlcircuit, and wherein the first current is temperature dependant; and acurrent minor that is coupled to the programmable current source and theboost circuit, wherein the current mirror receives a second current fromthe programmable current source.
 18. The apparatus of claim 17, whereinthe current minor further comprises a first current minor, and whereinthe boost current circuit further comprises: a second current mirrorthat is coupled to the first current mirror; a transition detector; aboost transistor that is controlled by the transition detector; and aswitch network that is coupled to the boost transistor, the secondcurrent minor, and the pair of differential terminals, wherein theswitch network couples the boost transistor and the second currentmirror to the pair of differential terminals during the voltagetransitions.
 19. The apparatus of claim 17, wherein the control circuitfurther comprises: a state machine; a variable impedance network that iscoupled to the programmable current source so as to receive the firstcurrent, that is coupled to the pair of differential terminals, and thatis controlled by the state machine; and a comparator that is coupled tothe variable impedance and the state machine.
 20. The apparatus of claim19, wherein the variable impedance network further comprises a DAC.